Discussing audio amplifier design -- BJT, discrete

[Jon Kirwan]

I'm no expert myself, but I'm willing to help where I can.


R2 provides both dc and ac feedback. DC for bias stabilisation
and setting the emitters of the output transistors to about half
of Vcc (more about that later).

This much I can see, immediately. And thanks for dealing
with more, later on.

For ac, it may be easier to think of it as current feedback.
Okay.

Q2 needs about +/-50uA peak of base
current at full drive. At signal frequencies, R2 (plus the much
smaller input impedance of Q1) is effectively in parallel with
the output.

R2 is connected from the output to an input, which
effectively doesn't move much after arriving at it's DC bias
point. As you later point out, the _AC_ input impedance is
lowish (near 600 ohms), so the 10k is pretty close to one of
the rails at AC, anyway. Is that a different way of saying
what you just said? Or would you modify it?

The output swings by about 4V peak at max power,
which has 400uA of negative feedback current going back through
R2. The input current requirement goes up by a factor of 9. IOW,
a negative feedback of 19db. This is substantially better than
nothing and should significantly reduce distortion and improve
frequency response.

Okay. This goes past me a little (as if maybe the earlier
point didn't.) I'd like to try and get a handle on it.

Let's start with the 4V peak swing at max power.

Since you are discussing AC and converting it 400uA current
via the 10k, I would normally take this to mean 4Vrms AC.
Which in Vp-p terms would be 2*SQRT(2) larger, or 11.3V which
I know is impossible without accounting for the BJTs, given
the 9V supply. So this forces me to think in terms of
something else. But what? Did you mean 4Vpeak, which would
be 8Vp-p? If so, that would be about 2.8Vrms. In that case,
wouldn't a better "understanding" come from then saying that
the negative feedback is closer to 280uA?

The next point is on your use of "goes up by a factor of 9."
Can you elaborate more on this topic? Where the 9 comes
from? For volts, not power, I think I can gather the point
that 20*log(9) = 19.085), so I'm not talking about that
conventional formula. I'm asking about the 9, itself, and
also your thinking along the lines of concluding that it
significantly reduces distortion. How does one decide how
much is enough?

The already low input impedance of Q1 is further reduced by the
negative feedback, so R1 represents practically the whole input
impedance of the amp. The -3db cutoff frequency is 1/2*pi*C1*R1
which is about 16Hz.

I think I follow. Leading towards a comment you make soon
below (about C3), C3's very low impedance even after beta
multiplication almost bypasses R3 completely so the impedance
is as you say, mostly depending upon C1 and R1. I think.

No. R4 is bypassed by C3 and has little effect on input impedance
except at very low frequencies.

Thanks for the knock in the head there. I had been ignoring
C3 for DC bais-point thinking and forgot to put it back in
when talking about AC loading. Your point is made.

It's mostly the internal dynamic emitter resistance that
determines Q1's input impedance. That resistance is 26/Ie at 20
deg C. Q2 is biased at about 7.7mA emitter current, giving about
3.4 ohms. Multiply that by hfe, add the ohmic base resistance and
you get Q1's basic input Z. I don't have my old data book handy,
but I think the AC126 had a typical hfe of about 150 and rbb of
maybe 100 ohms. This gives an input Z of about 600 ohms.

Okay. I get your point about small-case 're' based upon kT/q
and Ie. I'm mostly following here.

The output transistors need only about 0.1V each of Vbe to bias
them at a few mAs of Ic. D1 is germanium, but at the dc current
level flowing through it, two of them in series will have too
much voltage drop (I measured several samples). Ge transistors
have a more rounded knee than their Si counterparts in the Vbe
vs. Ic curve. So I felt that a single diode would present less
chance of thermal runaway for the output Trs and still cause a
reasonably low crossover distortion.
Thanks.

Oops. Have to go out for a while. Will take up the rest later.

Well, I think I'm roughly following so far. Please kick me
where I'm still off-track, if you feel you can afford the
moment to do it. I appreciate it very much.

Jon

 

[Jon Kirwan]

This is what happens without a booststrap: When it's Q5's turn to
conduct on the negative half-cycle of the signal, the base drive
current has to come via R6. At the same time, Q5's current is
pulling its emitter - and therefore the base - down towards
ground, decreasing the voltage drop across R6. This decreases the
base drive current available just when it's needed.

Now look at it modified with a boostrap:
http://img715.imageshack.us/img715/4259/boostrap.png

For simplicity, let R6 = R7. At steady-state, C will be charged
to about a quarter of Vcc. When Q5 pulls its emitter (and
therefore the positive electrode of C) towards ground, the
voltage across C cannot change instantaneously and will push its
negative terminal down too. Beyond a certain level of drive, the
junction of C, R6 and R7 will even go down past oV and become
negative with respect to ground. This maintains the voltage
across R6 at an approximately constant level.

Oops again. Guests this time. Will be back when I can.

I'm appreciating this very much. Printed the two and am
looking at both. I think I'm following, but need to sit down
and do a little paper calcs to make sure I burn it in a
little more. If I think of something useful, I'll post a
question or two. For now, I feel like I'm following you.

Jon

 

[Jon Kirwan]

I think this is where I left off earlier.


Your reasoning is correct. However, the output transistors Q2 and
Q3 need only about 100mV each of Vbe for Class AB bias. IIRC,
beta of Q1 is about 150 and Vbe at that level of current is about
0.12V.

Thanks for the adjustments. At least, I didn't wander too
far afield, for once. It's nice to hear I'm not totally out
of my depth.

I don't know about others, but with low voltage circuits, I
usually try to fix the quiescent voltage at the output mid-point
at slightly more than half of Vcc. This is because Q1's Ve plus
its Vcesat reduces the available downward swing of Q3's base.

Okay, that I follow.

For this design, I tentatively chose a target of 4.6V at Q3's
emitter. Add Q2's Vbe and that leaves 4.3V for R5 plus the
speaker's dc resistance. The speaker's resistance has only a
minor effect but, just for the heck of it, let's take it as 6
ohms. So Q1's Ic = 4.3/566 = 7.6mA.

I follow. Ve(Q2)=4.6V, Vbe(Q2)=0.1V, so Vb(Q2)=4.7V.
Assuming a solid 9V power source (not what a 9V battery is,
if that were used, but...), that leaves 9V-4.7V = 4.3V across
R5 and SPK1. Which sets a current to the Vb node of Q2 that
needs to be disposed of via D1 and then Q1. 7.6mA it is.

Q1's dc beta = 150,

That high? I had anticipated germaniums were lower. Okay.

so Ib is about 50uA,
Followed.

and Ie = 7.65mA = I(R4).

Yes. 7.6mA+0.05mA = 7.65mA that must be dumped into R4.
There will be some Q1 base current (another 50uA?) added to
Ie(Q1) that is ignored here. No problem. That would only
mean 7.7mA instead of 7.65mA for your calcs below.

7.65*33 places Q1's emitter at 252.45mV above ground.

or 7.7mA*33 = 254.1mV, if you add Q1's base drive?

That plus
Vbe of 0.12V gives Vb = 372.45mV.
Okay.

I(R3) = 372.45uA
I(R2) = I(R3) + Ib = 422.45uA
I(R2)*R2 = 4.2245V
V(R2) + Vb = (4.2245 + 0.37245)V = 4.59695V.

It just so happens that, in this case, common resistor values
produce almost exactly the desired quiescent bias level. If they
didn't, a slight departure from the target voltages would be
acceptable. In any case, tolerances on resistor values and
transistor characteristics could throw off actual values a bit.

I'm with you. Thanks for the care, here.

For such a simple design without a high level of audio quality as
the target, I wasn't too particular about the amount of signal
feedback as long as it's a reasonable amount. I chose a
compromise value for R3 first - low enough for bias stability so
that the current through it would be several times Ib, but not
too low to avoid excessive shunting of the signal input current.
Then I let the value of R2 be what it needs to be for correct
bias.

Okay. So I remember in an earlier post you talking about
400uA being a factor of 9 higher. Guessing that Q1's base
current is around the same area of 50uA, this would be a
factor of 8... not the 9 you mentioned before. But at least
I'm starting to see where that number 9 came from?

Then I calculate the amount of NFB as outlined in one of my
earlier replies and accept it if it's within reason. If I really
wanted more NFB, I'd parallel R2 with another resistor, but with
a capacitor in series to avoid upsetting the dc levels.

Got it. That would provide additional AC feedback but keep
the DC biasing. So you don't have to screw around balancing
R2 against two different considerations, you just "fix it"
with a patch like that. Makes total sense.

BTW, that
can be used to provide some bass boost by choosing the proper
values of cap and resistor.

Hmm... Lower frequencies would have less NFB, higher
frequencies more. Okay. There is also other areas where
higher frequencies are going to see less gain in this
design... Now I'm starting to wonder about phase shift not
exactly 180 degrees in the NFB over frequency. But I need to
sit down and think more.

That's a guesstimated value, partly empirical and partly based on
observation of other people's designs. No PCs and simulation
software 40 years ago. For such a simple circuit, I didn't bother
with complex calculations for loops and phase shifts that
wouldn't be precise anyway due to wide tolerances in component
characteristics.

The reason for the relatively high capacitance is that this was a
low-Z low-gain circuit. But I might have made a mistake in
showing it now as 2.2nF. I might have used something like 1nF.

Okay. Nuff said. I'll leave that for later thinking.

This is a variation of the bootstrap circuit I described in my
other post. R5 and the speaker serve the same functions as R6 and
R7 respectively in the other circuit.

Hmm. I generally get the thrust. I need to think more
closely about the value of it. But I'll take it as something
to explore more.

Jon

 

[Paul E. Schoen]

Have at me. I probably got a lot wrong in the above, but
that's my thinking exposed like a soft worm to be crushed. If
I learn in the process, crush away!

Have you used LTSpice (AKA SwitcherCAD)? Some time ago I tried some ideas
for linear amplifiers, and I came up with a design that is DC coupled and
uses just three MOSFETs. You can trim the quiescent current to trade
efficiency for crossover distortion. I originally made it with capacitor
coupling and a single supply, but I redid it with a dual supply and DC
coupling.

As a practical matter, the amplifier may be a bit unstable and prone to
overheating and even self-destruction if the bias current is set high
enough to eliminate crossover distortion. If you can tolerate a few percent
(tens of thousands of PPM, if you must), then it should be fairly stable.

This design is basically an output stage only, and has no voltage
amplification. That can be easily achieved with a simple class A input
stage.

When the output is driven just about to the rails, it puts out 5.5 watts
into 8 ohms, with input power of 8.9 watts, or 62% efficiency. I'm using
+/- 12 VDC rails and 7.7 VRMS input. There are two IRL3915 NMOS output
transistors (STD30NF06 also work), and one IRF7205 PMOS. I have not
actually built this circuit, and there are probably a number of problems
with making it work using actual components, but I think it's worth a try.
Of course, this is a MOSFET design and not BJT, but a similar circuit could
be built using BJTs if that is a requirement.

The ASCII file follows.

Paul

---------------------------------------------------------------

Version 4
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SYMATTR Value 10k
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SYMATTR InstName D1
SYMATTR Value 1N4148
SYMBOL diode 720 144 R0
SYMATTR InstName D2
SYMATTR Value 1N4148
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SYMATTR Value 1N4148
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SYMATTR Value 1N4148
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SYMATTR Value 220
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SYMATTR Value 220
TEXT 264 704 Left 0 !.tran .5
TEXT 416 704 Left 0 !;ac oct 5 20 20000

 

[Jon Kirwan]

Have you used LTSpice (AKA SwitcherCAD)?

Oh, yes. However, my interest is in _learning_. I use
LTspice to explore my own ignorance, at times. And I use it
to test what I think I've learned. But I don't "hack" with
it. Much.

Some time ago I tried some ideas
for linear amplifiers, and I came up with a design that is DC coupled and
uses just three MOSFETs. You can trim the quiescent current to trade
efficiency for crossover distortion. I originally made it with capacitor
coupling and a single supply, but I redid it with a dual supply and DC
coupling.

No FETs. I think I stated that in the beginning. There are
a variety of reasons why. But suffice it that I don't want
to go there... for now.

As a practical matter, the amplifier may be a bit unstable and prone to
overheating and even self-destruction if the bias current is set high
enough to eliminate crossover distortion. If you can tolerate a few percent
(tens of thousands of PPM, if you must), then it should be fairly stable.

This design is basically an output stage only, and has no voltage
amplification. That can be easily achieved with a simple class A input
stage.

When the output is driven just about to the rails, it puts out 5.5 watts
into 8 ohms, with input power of 8.9 watts, or 62% efficiency. I'm using
+/- 12 VDC rails and 7.7 VRMS input. There are two IRL3915 NMOS output
transistors (STD30NF06 also work), and one IRF7205 PMOS. I have not
actually built this circuit, and there are probably a number of problems
with making it work using actual components, but I think it's worth a try.
Of course, this is a MOSFET design and not BJT, but a similar circuit could
be built using BJTs if that is a requirement.

Yeah. That's a requirement. I've still some learning ahead
of me. But I'll take a look at the schematic and tuck it
away, at least. Thanks.

Jon

 

[Paul E. Schoen]

Oh, yes. However, my interest is in _learning_. I use
LTspice to explore my own ignorance, at times. And I use it
to test what I think I've learned. But I don't "hack" with
it. Much.


No FETs. I think I stated that in the beginning. There are
a variety of reasons why. But suffice it that I don't want
to go there... for now.


Yeah. That's a requirement. I've still some learning ahead
of me. But I'll take a look at the schematic and tuck it
away, at least. Thanks.

OK. So I made a similar amplifier output stage using two 2N3055s, and a
2N3904 and a 2N3906. With 8.4 VRMS input the output is 7.3 VRMS into 8 ohms
for 6.66 watts. Input power is 9.97 watts, efficiency is 67%. Some very
slight crossover distortion. 6 mA drive current (about 1.2k input
impedance). Looks good 20 Hz to 20 kHz. I added an output inductor which
affects output at higher frequencies. LTSpice ASCII follows.

Paul

-----------------------------------------------------------

Version 4
SHEET 1 1304 744
WIRE 736 -176 192 -176
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WIRE 976 -176 864 -176
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TEXT 264 704 Left 0 !.tran .5
TEXT 416 704 Left 0 !;ac oct 5 20 20000

 

[Jon Kirwan]

<snip>
OK. So I made a similar amplifier output stage using two 2N3055s, and a
2N3904 and a 2N3906. With 8.4 VRMS input the output is 7.3 VRMS into 8 ohms
for 6.66 watts. Input power is 9.97 watts, efficiency is 67%. Some very
slight crossover distortion. 6 mA drive current (about 1.2k input
impedance). Looks good 20 Hz to 20 kHz. I added an output inductor which
affects output at higher frequencies. LTSpice ASCII follows.
<snip>

Got it and saved it. Sziklai pair on one side, Darlington on
the other. 3 NPN, 1 PNP. Now I'm thinking about tubes (they
don't come in complementary form) and using a signal splitter
and NPNs "all the way down!"

Jon

 

[pimpom]

R2 is connected from the output to an input, which
effectively doesn't move much after arriving at it's DC bias
point. As you later point out, the _AC_ input impedance is
lowish (near 600 ohms), so the 10k is pretty close to one of
the rails at AC, anyway. Is that a different way of saying
what you just said? Or would you modify it?

That's another way of putting it, yes.

Okay. This goes past me a little (as if maybe the earlier
point didn't.) I'd like to try and get a handle on it.

Let's start with the 4V peak swing at max power.

Since you are discussing AC and converting it 400uA current
via the 10k, I would normally take this to mean 4Vrms AC.
Which in Vp-p terms would be 2*SQRT(2) larger, or 11.3V which
I know is impossible without accounting for the BJTs, given
the 9V supply. So this forces me to think in terms of
something else. But what? Did you mean 4Vpeak, which would
be 8Vp-p? If so, that would be about 2.8Vrms.

Yes. It's 4Vp, 8Vp-p and 2.8Vrms. I wanted to give you a mental
picture of how much the output voltage can swing. Each output Q
has about 4.4V of Vce available, and about 4V before hard
saturation is reached (these are all round figure values). That's
4V peak for a sinusoidal wave form.

In that case,
wouldn't a better "understanding" come from then saying that
the negative feedback is closer to 280uA?

Yes, it's 280uA rms. But I was talking in terms of the maximum
amplitude of instantaneous change, which is why I used the terms
"swing" and "peak".

The next point is on your use of "goes up by a factor of 9."
Can you elaborate more on this topic? Where the 9 comes
from? For volts, not power, I think I can gather the point
that 20*log(9) = 19.085), so I'm not talking about that
conventional formula. I'm asking about the 9, itself, and

Without feedback, the input transistor Q1 needs 50uA of AC input
signal to drive the output Qs to full power output (still talking
in terms of peak to avoid confusion). With NFB, we need an
additional 400uA to overcome the current fed back from the
output. That's a total of 450uA peak, which is 9 times the
original 50uA.

Actually, I made an error when I cited the 50uA figure. Q1 is
biased at Ic = 7.6mA, Ib = 50uA. But only 5mA peak is needed from
Q1's collector to drive the output transistors. Divide that by
Q1's hfe of 150 and you get 33uA (peak) of AC signal current
needed into the base of Q1. The corrected total needed from the
signal source is now 433uA. The gain reduction factor due to NFB
is now 13 instead of 9. That's 22db (feedback is usually given in
db).

also your thinking along the lines of concluding that it
significantly reduces distortion.

The basic principle of NFB is that it reduces THD and extends
frequency response by a factor equal to the feedback ratio. So,
in our example, if you have 10% THD without feedback, it will
drop to 0.77% with the feedback factor of 13. But there are
caveats. E.g., phase shifts can cause undesireable effects,
especially with large amounts of feedback. I'm afraid a detailed
treatment of such things is really outside the scope of this
discussion - unless someone else is willing to take it up.

How does one decide how much is enough?

For one thing, how much distortion one is willing to put up with.
Another factor is input sensitivity, or IOW, how much gain is
needed. E.g., to drive the 1W amp to full output, we need 433uA
peak (306uA rms) from the signal source into 1k. That's 306mV
rms, plus some millivolts at the b-e junction. Say about 0.32V
rms total input voltage into about 1k input impedance.

To present the basic concepts, I've made several approximations.
E.g., I neglected the shunting effect of R2. Besides, the input
resistance of Q1 is constant at 600 ohms only for very small
signal amplitudes relative to the quiescent dc levels. This
dynamic input resistance changes significantly with large signal
swings and adds distortion while also complicating precise
calculations.

 

[pimpom]

On Jan 27, 9:51 pm, Jon Kirwan <[email protected]>
wrote:

"I'd probably replace the two diodes with
one of those BJT and a few resistor constructions I can't
remember the name of (which allows me to adjust the drop.)"

First Jon I know less about amplifier design than you do...
That said,
I would be careful about replacing the diodes in the push-pull
stage.
Way back in college I had a Sony stero amp that I had to fix.
It came
with a nice circuit diagram. I seem to recall that the bias
diodes
in the push pull stage were thermally attached to the same heat
sink
that held the output transistors. As the output transistors
warm up
their Vbe drop decreases. You want the bias diodes to track
this
change. Or else the whole thing could 'run-away' on you. ...
degenerative emmiter resistors (as you suggest) will help some.

I like the biasing scheme mentioned by Jon and use it for all my
designs except the early ones using germanium transistors, though
I don't know the name either. The biasing transistor can be
mounted on the output transistors' heatsink for temperature
tracking.

I like it because it's versatile and a single transistor can be
used to bias several transistors with their b-e junctions in
series as long as they are mounted on a common heatsink.
http://img691.imageshack.us/img691/2075/bias.png

My personal preference is to place the bias adjustment pot R3 in
this position rather than with R1. It ensures that any accidental
loss of contact by the pot's wiper arm will reduce the total bias
whereas placing it with R1 will have the opposite effect and
could cause excessive quiescent current in the output
transistors, possibly getting them to overheat.

 

[Jon Kirwan]

Hi Jon, I'm enjoying your posts.

Thanks. I feel like I'm way behind some curves, but it's fun
taking a moment to think about things and it is fantastic
that anyone else is willing to help talk about things with
me. That is priceless. So the real thanks go to those who
are sharing their knowledge and experience here.

What's a pin driver?

Hmm. I think I first heard the idea when talking about
testing ICs, to be honest. But imagine instead a micro with
software to test some discrete part (could be an IC, too,
that that's more complex.) For example, to automatically
derive some modeling parameters for a BJT.

Take a look at this datasheet, for an example of the features
one might support:

http://www.analog.com/static/imported-files/Data_Sheets/AD53040.pdf

I made a nice
switchable current source (10nA to 1mA) from a voltage reference,
opamp and switchable resistors. (circuit cribbed from AoE.)

I'd require at least one that can either sink _or_ source to
the pin. And that would be only one of the pin driver's
required features. I think the datasheet mentioned above
provides some more. But that part is expensive and not
readily available to us hobbyist types and doesn't teach me
anything about various trade-offs I might want to make or how
to design it at all, besides.

Jon

 

[Jon Kirwan]

Sorry Jon,

I'm stuck on google groups for a little while - I can't believe people
actually use it full time or that google could make an interface this
bad. (I suspect it is very fine for simple threads) anyway...

Cripes. Google didn't even show the thread when I'd looked,
a day ago or so. And it had been around for at least 24
hours by then. Used to be the case that google groups would
show the posts within an hour or so. Doesn't seem to be
true, anymore. If not, there is no possibility of having a
discussion very quickly via google. It would greatly
lengthen out the interactions. Maybe that's on purpose, now,
to cause people to find some other solution?

Sorry.

Don't be. I was just explaining myself, not complaining
about your usage.

A little experience will get you into the right ballpark when
estimating what you could expect for distortion. It is basically the
same "rules" as you would see with op-amps - the more linear it is to
start with the better. Higher bandwidth stages generally mean you can
use more negative feedback to eliminate distortion - but the lower the
final gain the more instability is likely to become a problem. And bad
circuit layout can increase distortion (and even more so hum and
noise) easily by a factor of 10.

As for how low you need distortion to be one rule of thumb (I forget
the reference) is to be clearly audible the message must be 20db above
the background noise and to be inaudible distortion has to be 20db
below the background noise - which pretty much sets "low" distortion
for PA and similar uses at 1% or 10000 ppm. For HiFi the "message" has
a high dynamic range and you (allegedly) want a distortion figure at
least 20db below that. So a 60 db signal range 0.0001% (or 100PPM).
The you start getting into all kinds of trouble with power output /
dynamic range of the amp etc and you relies that it is all a
compromise anyway. You do the best you can within the restrictions of
the job description.
Understood.


Yes and No. All the published circuits are made by people who want to
sell transistors,

A concern I care not the least about. My _real_ preference,
were I to impose it on the design, would be to use ONLY
PN2222A BJTs for all the active devices. One part. That's
it. Why? Because I've got thousands of them.

Literally. Something like 22,000 of the bastards. I give
them away like popcorn to students at schools. Got them
_very cheaply_. So if I were pushing something, I'd be
pushing a 10W PN2222A design, use signal splitting approach
probably (because it's the only way I think think of, right
now), and distribute the dissipation across lots and lots of
the things.

What to go there?

not audio systems, power supplies or transformers.

Got it.

As a result the power supply is often assumed to be regulated, which
is not true in this case, or the power supply is treated in a very
perfunctory manner that is not at all compatible with good design.

In this case you have the voltage you need for the 10 watts, plus
voltage drop for the driver circuitry and output stage , plus ripple
voltage, plus whatever is required for transformer regulation and
mains regulation. When you add it all up you might find that a chosen
transistor/component is actually not at all suitable for the job. Back
to the drawing board. Change this change that recheck everything again
etc.

In this case, though, there is nothing particularly
remarkable about the rails. Taken across the entire span,
even, doesn't exceed the maximum Vce of a great many BJTs. So
no real worry there. But I see some of where problems may
arise. Luckily, at this level I can side-step worrying about
that part and get back to learning about amplifier design,
yes?

If you do the power supply first you have the figures needed for your
worst case already. It saves time and makes a better result (no
tendency to comprimise to save all the calculations already done).

Well, does this mean we should hack out the power supply
first? I'm perfectly fine with that and can get back to you
with a suggested circuit and parts list if you want to start
there. We could settle that part before going anywhere else
and I'd be happy with that approach, too, because to be
honest I don't imagine it to put a horrible delay into
getting back to amplifier design. So I'm good either way.

I wasn't going to prompt, but it is close to the sort of thing, I
think, you should be aiming for . As someone has already noted (I
would attribute you if I wasn't on GG, I'm sorry) it has been drawn up
for a single supply, rather than a more common (for this size /
configuration) split supply.

I had assumed we'd be using a split supply.

I had assumed a speaker would be hooked up via a cap to the
output, so DC currents into a speaker coil would be removed
from any concern. But I was also holding in the back of my
mind the idea of tweaking out DC bias via the speaker and
removing the coupling cap as an experiment to try. And if
so, I'd pretty much want the ground as a "third rail."
(Playing just a bit upon the Chicago parlance about the once
dangerous rail in their transit system.)

(I don't really care too much about arguing about

Correct. Theory says it does nothing. I practice the theory but have
the occasional heretical belief about that.

Actually, I think I've read that theory says it is _better_
to be removed. The reason seemed pretty basic, as it's
easier to get close to a balanced current split; and this, I
gather, lowers 2nd harmonic distortions produced in the pair
-- notable more on the high frequency end I suppose because
gain used for linearizing feedback up there is diminishing
and can't compensate it.

In other words, it's not neutral. It's considered to be
better if I gathered the details. Then even better, the
current mirror enforces the whole deal and you've got about
the best to be had.

Of course, mostly just being a reader means I have no idea
which end is up. So I might have all this wrong.

and better still both R3 and R4 should be

This would provide more differential gain.

_and_ improve distortion because the currents are forced to
be balanced in the pair, yes?

In the right configuration it would reduce the common mode signal gain
of things like mains hum and supply ripple (you mentioned power supply
isolation before).

Yes, that's how I thought about it.

Also, from another (what do you call it branch? thread?) you were
discussing boot-strapping R6. This is not done so much as amplifiers
get bigger but a BJT configured in the same way as the replacement for
R5 is very common. I'm leaving the details to you - perhaps there is a
way to reduce component count without affecting performance. (I am
hoping this is what you wanted "nutting it out for yourself")

Yes! I don't want things handed on a platter. But I also
don't want to have to rediscover all of the ideas by making
all of the mistakes, either. This is the kind of "pointer"
towards something that I like a lot. It gives me a place to
think about something, but leaves me some reason to have to
do so and that helps me own it better.

One general truth about learning is that you don't present
someone with a problem so out of their depth that they have
no chance at it. Doing that means they fail, they feel like
a failure, and it causes a student to just want to go away.
They lose motivation, usually, in cases like that. On the
other hand, providing no difficulty at all merely means
repetition of what they already know and they grow bored from
that, too. Finding the sweet spot where a student is faced
with interesting problems that are not already known, but
perhaps within reach of grasping at with some effort, is the
key. Then it can be fun, educational, and motivate.

That's what you just did for me.

I assume the input impedance of that example

Okay.

Yes, but you should probably think of a whole passive network to
filter out low and high frequency - (think what happens if you amp is
operated near a source of RF)

Well, every trace picks up like little antennae. All kinds
of trace voltages appearing here and there. Not good.

So. Can you make an audio amplifier that can withstand a
microwave oven environment and deliver good performance while
irradiated with 1kW banging around in there?

Depending on transistors layout etc it might not be needed, but more
often it is the size that is the question.

I was thinking it helped locally linearize the VAS section
and that such would be "good" most anywhere. But I am just
taking things without having worked through them on my own.
So...

Vbe multiplier...

Okay. Thanks.

Why would/should you use them?

I'm still thinking about that. In general, I was thinking
about them because of the "little re" that is kT/q based in
each BJT, and varies on Ie. Since Ie is varying around, I
was thinking about something fixed there to overwhelm it and
"make it knowable" for the design, I suppose. Maybe that's
all wet, given your query. I'll toss the idea off the side,
for now.

Not at all.

Thanks for that. I'm just glad to be able to talk to someone
about any of this, at all. So please accept my thanks for
the moments you are offering.

Is there a way you could post a schematic of where your thinking is
and what you would like to discuss - there is no need for a complete
circuit.

Yes. I can use ASCII here, for example. But before I go off
into the wild blue with this, do you want to focus on the
power supply first? Or just jump in on the amplifier?

Jon

 

[Jon Kirwan]

A concern I care not the least about. My _real_ preference,
were I to impose it on the design, would be to use ONLY
PN2222A BJTs for all the active devices. One part. That's
it. Why? Because I've got thousands of them.

Literally. Something like 22,000 of the bastards. I give
them away like popcorn to students at schools. Got them
_very cheaply_. So if I were pushing something, I'd be
pushing a 10W PN2222A design, use signal splitting approach
probably (because it's the only way I think think of, right
now), and distribute the dissipation across lots and lots of
the things.

What to go there?

I meant "Want to go there?"

One part of the brain says "type out word for concept X on
the keyboard," and that gets passed down to a low-level
manager function which maps concept to English word, gets the
wrong hash bucket ID for a "nearby word" and the passes on
motor instructions for mirror neurons driving the hand which
then types the wrong word. Meanwhile, eye reads "want" text,
this gets translated into a concept which is promptly ignored
because the concept already resides in local cache storage
and doesn't need replacement. So the brain checks that what
I wrote is what I intended, glancing quickly over it and gets
the cached version and matches everything up nicely and moves
on.

hehe. This is either a hardware problem or a software
problem, depending upon point of view...

Jon

 

[Jon Kirwan]

eye reads "want" text

hehe. Or "What" text....

Nesting all the way down...

Jon

 

[Jon Kirwan]

I like the biasing scheme mentioned by Jon and use it for all my
designs except the early ones using germanium transistors, though
I don't know the name either. The biasing transistor can be
mounted on the output transistors' heatsink for temperature
tracking.

I like it because it's versatile and a single transistor can be
used to bias several transistors with their b-e junctions in
series as long as they are mounted on a common heatsink.
http://img691.imageshack.us/img691/2075/bias.png

Yeah. That's what I was thinking about. Just some thought
here to add. As you carefully point out, tying it to the
heat sink of the output BJTs to hold them near each others'
temperatures seems important. Vbe varies a great deal over
temperature.

The voltage across this structure is:

Vbe * (1 + R1/(R2+R3))

However, it's also true that the output BJTs are also
experiencing similar (but _not_ the exact same as they aren't
necessarily even from the same manufacturer or family)
changes in Vbe. So it's actually a kind of "good thing" to
have the voltage held between the output BJT bases vary as
the output BJTs temperatures vary.

Question is, is a random selection of a BJT for this purpose
okay? Or does it need to be carefully considered, taken
together with the output BJT characteristics? It seems to me
that some care is needed here, even assuming good temperature
coupling occurs.

Also, I think I've seen some examples where there is a
collector resistor added to this structure, with Q2's base
kept tied directly to Q1's collector lead. What is the
reasoning here? (I believe in the cases I saw, there was a
current source [not a resistor] feeding at the top. I
started to work the equations to show the relationship, but
then realized that there is also base current drive to the
upper side of the output transistors involved and then
decided to just ask, instead of wandering all over the place
right now.)

My personal preference is to place the bias adjustment pot R3 in
this position rather than with R1. It ensures that any accidental
loss of contact by the pot's wiper arm will reduce the total bias
whereas placing it with R1 will have the opposite effect and
could cause excessive quiescent current in the output
transistors, possibly getting them to overheat.

Makes sense.

Thanks,
Jon

 

[Paul E. Schoen]

A concern I care not the least about. My _real_ preference,
were I to impose it on the design, would be to use ONLY
PN2222A BJTs for all the active devices. One part. That's
it. Why? Because I've got thousands of them.

Literally. Something like 22,000 of the bastards. I give
them away like popcorn to students at schools. Got them
_very cheaply_. So if I were pushing something, I'd be
pushing a 10W PN2222A design, use signal splitting approach
probably (because it's the only way I think think of, right
now), and distribute the dissipation across lots and lots of
the things.

I design a lot of things that way as well. I have (hundreds of) thousands
of parts that I got about 20 years ago, and I really like to use them
wherever possible. You can check my website where I have some of these
parts listed as surplus sales and I'd really like to get rid of them where
they might be used rather than hauling them to the dump. Take a look and if
you can use anything I'll see if it's worthwhile to send them to you for
little more than the cost of shipping (probably USPS flat rate). My website
is www.pstech-inc.com, and just look for the link to surplus parts. If that
doesn't work, try http://www.smart.net/~pstech/surplus.htm and
http://www.smart.net/~pstech/PARTS.txt and
http://www.smart.net/~pstech/PARTS.xls.

I have a lot of MPSA06 NPN transistors, so I use them wherever possible. I
also have a few thousand MJE170 PNP Power transistors (40V, 3A, 12W). And
lots of 2N6312 in TO-66 metal cans (PNP 40V 5A 75W) and about 600
Thermalloy 6060 heat sinks that can be used for them, as well as other case
types. If you like SCRs I have about 600 of 2N6504 which is 35A at 50 V.

If you need a transformer for a power supply I have a couple hundred Signal
241-6-16 which can be used to make a raw +/- 8-10 VDC supply and with a few
more capacitors and diodes makes a nice +/- 20 VDC supply at about 1 amp.
And I have an armload of capacitors such as 500 uF 50V, 1500 uF 50V, and
even some 4500 uF 50V in big blue metal cans. And a few handsful of 1N4003
and 1N4004 rectifiers.

If you could come to my place near Baltimore, MD I could give you a
"shopping spree" where you could fill a few bags and boxes with all sorts
of goodies. Lately I am realizing that almost any new design I do will be
with SMT components and newer parts, and there are only a few one-off
projects that I might make using these older components. Some of them have
been stores so long in a damp, unheated building that the leads are
difficult to solder, and some resistors have actually soaked up enough
moisture to change value. (That is what a friend told me, and he also said
they were restored to normal by baking them for a while).

I've sent "care packages" to others in the past. I don't expect to make any
money selling/giving away these parts but I just want to be compensated for
shipping cost. I don't know if I have some of these parts and if I do I
might not even be able to find them, but I think I can supply enough parts
for you to build a good amplifier and other projects.

Paul

 

[Paul E. Schoen]

Makes sense.

As a quick precaution against thermal runaway, a thermistor from base to
emitter, and thermally tied to the case, should shut off base drive if
things get too hot. Or just put a thermal switch on the heat sinks and use
it to shut off the supply to the whole shebang.

Paul

 

[Jon Kirwan]

<snip>

If you could come to my place near Baltimore, MD I could give you a
"shopping spree" where you could fill a few bags and boxes with all sorts
of goodies. Lately I am realizing that almost any new design I do will be
with SMT components and newer parts, and there are only a few one-off
projects that I might make using these older components. Some of them have
been stores so long in a damp, unheated building that the leads are
difficult to solder, and some resistors have actually soaked up enough
moisture to change value. (That is what a friend told me, and he also said
they were restored to normal by baking them for a while).

I've sent "care packages" to others in the past. I don't expect to make any
money selling/giving away these parts but I just want to be compensated for
shipping cost. I don't know if I have some of these parts and if I do I
might not even be able to find them, but I think I can supply enough parts
for you to build a good amplifier and other projects.

Paul

Paul, I'll write under separate cover, directly. There are
some thoughts I'd like to explore more, if that's okay. I
can also provide a 501(c)3 for tax purposes, as well as cash
compensation. That may also help a little. But we can talk
about that off-line.

Jon

 

[Phil Allison]

"John Larkin is lying IDIOT

The classic output stage biasing scheme uses small emitter resistors
and biases the output transistors to idle current using a couple of
junction drops between the bases, or a Vbe multiplier with a pot. Both
are good ways to have a poorly defined idle current and maybe fry
transistors.

** Done correctly, either way produces a stable bias situation in the output
stage.

Larkin has no idea how it is done - cos Larkin is bullshitting asshole.

1. Use zero bias. Connect the complementary output transistors
base-to-base, emitter-to-emitter. Add a resistor from their bases to
their emitters, namely the output. At low levels, the driver stage
drives the load through this resistor. At high levels, the output
transistors turn on and take over.

** Guarantees serious x-over distortion.

Zero bias can be done, but never so crudely as that.

2. Do the clasic diode or Vbe multiplier bias, but use big emitter
resistors. Parallel the emitter resistors with diodes.

** No need to ever use emitter resistors of more than 1 ohm.

With the usual 0.33 to 0.47 ohm resistors, parallel diodes have barely any
effect.

Very few power amp designs have ever used them - SAE brand amps from the
late 1970s being one exception.

In both cses, the thing will be absolutely free frfom thermal runaway
issues and won't need adjustments.

** Vbe multipliers always need adjustment to suit the actual devices in use.

Both need negative feeback to kill crossover distortion.

** One thing that NFB is notoriously very poor at doing.



..... Phil

 

[Paul E. Schoen]

Discrete-transistor audio design, being such an ancient practice,
tends to refer to history and authority rather than design from
engineering fundamentals.

If I were designing an audio amp nowadays (which I certainly aren't)
I'd use mosfets with an opamp gate driver per fet. That turns the fets
into almost-perfect, temperature-independent, absolutely identical
gain elements. That's what I do in my MRI gradient amps, whose noise
and distortion are measured in PPMs.

ftp://jjlarkin.lmi.net/Amp.jpg

Why keep repeating a 50-year-old topology when you could have a little
fun?

Nice photo. But how about a schematic to show how you implemented the
design? I made a simulation of an amplifier design where I used a MOSFET
output stage similar to my other post, and I closed the loop with a single
op-amp and appropriate negative feedback.

I think the OP is looking for a basic learning experience using the
simplest components. It may be argued that a MOSFET is simpler than a BJT,
but experience with both is a good idea. If the object were just to make an
audio amp, there are single package designs and kits that will do the job
nicely.

I prefer making the circuit using LTSpice, but it can be a thrill to build
something with real components. There are usually some gotchas that cause
unwanted behavior not indicated in the simulation. But I need a real reason
to build something, other than practice with soldering and handling
components, so I rarely commit these designs to copper and silicon.

Paul

 

[Jon Kirwan]

That's another way of putting it, yes.

Okay. Thanks.

Yes. It's 4Vp, 8Vp-p and 2.8Vrms. I wanted to give you a mental
picture of how much the output voltage can swing. Each output Q
has about 4.4V of Vce available, and about 4V before hard
saturation is reached (these are all round figure values). That's
4V peak for a sinusoidal wave form.

Got it.

Yes, it's 280uA rms. But I was talking in terms of the maximum
amplitude of instantaneous change, which is why I used the terms
"swing" and "peak".
Understood.


Without feedback, the input transistor Q1 needs 50uA of AC input
signal to drive the output Qs to full power output (still talking
in terms of peak to avoid confusion). With NFB, we need an
additional 400uA to overcome the current fed back from the
output. That's a total of 450uA peak, which is 9 times the
original 50uA.

Okay. Let me put it in my own words. Since you chose to
stick a 10k in for the NFB, and since it supplies a peak of
400uA into the input node and since Q1 itself requires its
own 50uA (corrected below), then the signal source itself
must "comply" by supplying 450uA peak into the input node.
And that is where you got your 9. If that is it, I've got
it.

Actually, I made an error when I cited the 50uA figure. Q1 is
biased at Ic = 7.6mA, Ib = 50uA. But only 5mA peak is needed from
Q1's collector to drive the output transistors. Divide that by
Q1's hfe of 150 and you get 33uA (peak) of AC signal current
needed into the base of Q1. The corrected total needed from the
signal source is now 433uA. The gain reduction factor due to NFB
is now 13 instead of 9. That's 22db (feedback is usually given in
db).

Okay. I can compute the numbers. I think the problem I'm
having with this, as a mental concept, is that the feedback
is _from_ the complementary BJT outputs at their emitters
_backwards_ into the node at the base of Q1. Not outwards
from that node _towards_ the emitters.

Let me think a little about this from an AC point of view,
not DC. The output is supposedly running with about 4Vpeak
into an AC divider made up of what you've earlier described
as about 600 ohms to ground via Q1 and about 1k for R1. So
given that, we are talking about 1k from input to base node,
600 ohms from base node to ground, and 10k from 4V peak
output into base node. The 4V peak has the opposite sign as
the input because of the relationship of Q1's collector to
its base voltage. I'm just spouting things here without a
lot of understanding, so bear with me.

The change in Ic of Q1 for a change in the base voltage
(computing a transconductance of Q1) is 1 divided by re,
which you computed as 3.4 ohms earlier. So gm=294mSeimens,
assuming the emitter (small signal wise) follows the base
exactly. A 1mV input change at the base yields 1mV*294mS or
294uA change in the collector, ignoring the 100 ohm rbb you
earlier mentioned for now. Multiplied by the collector load
of about 566 ohms (your figure) produces 166mV change at the
collector. A voltage gain (base node to output) of -166.

This 166mV change is fed back via the 10k into an existing
divider composed of the 1k and the beta-multiplied 3.4 ohms
to ground (if I'm following you.) So with the rbb of 100,
about 600 ohms as you mentioned (still AC-minded.)

So back to the 1k from input, 600 to ground, 10k in from that
-166mV change in response to a +1mV change at the base node.
The 1k/600 divider means the input had to vary by about
2.66mV to achieve that node change. That means our voltage
gain wasn't really -166, but more like 62.5 -- audio input to
Q1 collector and then to output drive.

Am I going around the barn about right, so far? Here's what
worries me now. The postulated thevenin base node change of
1mV is through a thevenin of about 375 ohms (the 1k and 600
ohm splitter.) The 10k feeds into this from the other side
with -166mV there. If I imagine 1mV on one side via 375 ohms
and -166mV on the other side via 10k ohms, what is the node
itself at? Well, (1mV*10k+-166mv*375)/(10k+375) is -5mV or
so. This is a lot, isn't it? And it is more than enough to
oppose the postulated thevinin change at the input node of
+1mV.

So I follow the calculation of 22db. The problem I'm having
is with what kind of signal will result at Q1's collector.

Okay. Granted. I am sure my reasoning fails on some points
you will make clearer. I'm just trying to see this in a
variety of ways rather than just let you tell me stuff
without running through different thinking to see if I get to
the same place. So what did I do wrong here? I can't argue
with success and I know I'm not doing this right. But there
it is.

The basic principle of NFB is that it reduces THD and extends
frequency response by a factor equal to the feedback ratio. So,
in our example, if you have 10% THD without feedback, it will
drop to 0.77% with the feedback factor of 13. But there are
caveats. E.g., phase shifts can cause undesireable effects,
especially with large amounts of feedback. I'm afraid a detailed
treatment of such things is really outside the scope of this
discussion - unless someone else is willing to take it up.

I can do phase shift calcs given simple cases and if I take
into account all the necessary parts (which, being ignorant
about all this, I'm unlikely to do without more thought
experiments to clarify my thinking.) So when I get to that
point where I can actually walk myself along better, I'll be
able to handle that (I hope.)

For one thing, how much distortion one is willing to put up with.
Another factor is input sensitivity, or IOW, how much gain is
needed. E.g., to drive the 1W amp to full output, we need 433uA
peak (306uA rms) from the signal source into 1k. That's 306mV
rms, plus some millivolts at the b-e junction. Say about 0.32V
rms total input voltage into about 1k input impedance.

To present the basic concepts, I've made several approximations.
E.g., I neglected the shunting effect of R2. Besides, the input
resistance of Q1 is constant at 600 ohms only for very small
signal amplitudes relative to the quiescent dc levels. This
dynamic input resistance changes significantly with large signal
swings and adds distortion while also complicating precise
calculations.

Okay. I'm going to leave things with the above "issue" laid
out for you. It's bugging me right now.

I am refusing, by the way, to attempt any simulation. I
don't want to be "told" something by a simulator or handed
things on some platter. I want to try and work through my
thinking, find the flaws, slap myself for them, get back and
try again, until I'm good from a paper-and-brain point of
view. _Then_ I'll go and check it out to see where the chips
fall in the simulator. Might bring up a good question at
that point. But until I can get the gross aspects down, that
won't really matter.

Thanks so much for your efforts so far. It's been helpful to
me, at least.

Jon